1. Field of the Invention
This invention pertains to the field of digital signal processing and, in particular, to the field of analog to digital (A/D) conversion.
2. Background Art
In most digital video processing, eight bits are used to represent each sample. This number has been found adequate to reduce the effects of quantizing to an acceptable level when the signals being digitally coded are composite video signals that have already been gamma corrected. As compared with the conversion of composite PAL or NTSC signals into digital form, however, the A/D conversion of color component signals prior to video processing (especially gamma correction) requires a significantly greater bit accuracy. More particularly, when signals are digitally coded before gamma correction, the visibility of quantization noise is increased in low light. (This is because gamma correction increases the gain for signals near black.) Low level component signals therefore need to be coded to more than the usual eight bits coding resolution.
Coding requirements for a telecine (a machine for converting film images into video signals) are relatively more stringent than for other video systems, e.g., a video camera. This is because the film scanned by the telecine has its own characteristics which have to be compensated in order to provide good television pictures. In particular, the telecine must process a much larger input contrast range than a live camera because of the expansion in contrast produced by the film gamma. As a result, correction down to much lower values of gamma is required from a telecine in order to compensate for both the display tube gamma and the film gamma. In addition to gamma correction, film exposure may need to be compensated and the opertor's preference for color and density correction may need to be allowed for.
If the digitization of the primary R, G, B signals take place before gamma correction, at least 11 bits are ordinarily needed for a telecine video processing channel. High speed, single-stage A/D converters (e.g., parallel, or flash, converters) with such extended resolution or multi-state A/D converters (e.g.., subranging converters), however, either are not currently available or are inconvenient to implement (cost, availability, etc.). However, the extra resolution is needed only when the input signal is small. To achieve greater resolution without the greatly increased complexity of developing an extended range A/D converter, 8-bit A/D converters have been modified so as to insert analog preamplification whenever the input signal falls below a predetermined threshold. The preamplification factor is ordinarily an exact power of 2; thus the eight-bit word from the A/D converter can be located within a longer word by simply displacing it by the appropriate number of binary places.
Tests of visibility of quantization effects on signals that have been linearly coded and subsequently gamma corrected show that the smallest fractional change in perceived luminance that can be seen is about 2%, and the perceived fractional change does not start to rise rapidly above 2% until the signal level falls below about 15%, a preamplification factor of 8 can be applied to input signals falling below 12.5% of peak (the predetermined threshold). This will give three extra bits at signal levels below 12.5% (see "A Digital Telecine Processing Channel," by A. Oliphant and M. Weston, SMPTE Journal. July 1979, vol. 88, pp. 474-480). For example, the model B3410 Telecine, manufactured by Marconi Communications Systems Ltd., England, incorporates 11 bit A/D conversion provided by an 8 bit A/D converter and a gain switching system. The A/D conversion has a normal accuracy range and a fine accuracy range. Input signals falling between 12.5% and 100% of peak white are digitized over the whole 8 bit range of the converter to provide the eight MSBs of the output signal. Input signals falling below 12.5% of peak are amplified eight times and are then digitized to provide eight LSBs of the 11 bit output signal. The output signal is provided as shown below for high contrast film:
______________________________________ bit 10 9 8 7 6 5 4 3 2 1 0 ______________________________________ fine accuracy O O O X X X X X X X X (&lt;1/8 max) normal accuracy X X X X X X X X O O O (&gt;1/8 max.) ______________________________________
(see "Digital Video Processing for Telecine," by R. Matchell, IBC 1981, IEE Conference Publication N110, IEE London, U.K., pgs. 41-45; "The Marconi B3410 Line Array Telecine," by R. Matchell, SMPTE Journal, Nov. 1982, pp. 1066-1070).
Such level-dependent A/D converters are sometimes called dual range (or range-changing) converters, and typically include two separate conversion paths, one path with a multiple of the gain of the other path. A comparator, or a comparison-type operation, switches the input samples from one path to the other as the input video signal level passes a preset threshold level. One approach is to "gang" together two flash A/D converters such that the low gain path in effect has a coarser step size than the high gain path (see the dual-ranging A/D converter disclosed in "High-resolution digitization of photographic images with an area charge-coupled device (CCD) imager" by J. R. Milch, SPIE Vol. 697, Applications of Digital Image Processing IX (1986), pp. 96-104). A known dual range A/D converter of this type is shown in FIG. 1. Input analog signals are applied to a first M bit A/D converter 10 that is used in a low gain signal path A to generate first digital signals and a second M bit A/D converter 12 that is used in a high gain signal path B to generate second digital signals. The low gain path A is provided with unity gain from an amplifier 14 while the high gain path B is provided with a gain of 2**N from an amplifier 16. The low gain path is used when the input signal V.sub.IN is greater than (Vmax/2**N) and the high gain path is used when the input signal is less than or equal to (Vmax/2**N). (Vmax is the maximum value that signal V.sub.IN may become.) A comparator 18 compares the code values of the second digital signals to a suitable switch-over code word (usually just less than Vmax/2**N) and thereby controls a digital multiplexer 20 to select the data path to be utilized. (An overflow flag (OVF) from the second A/D converter 12 could also be used to control the multiplexer 20. Similarly, an analog comparator could be used to compare the analog input to the A/D converters 10 and 12.)
The multiplexer 20 includes registers 20a and 20b for assembling an output code word from the M data bits and N zero bits, the latter being joined to the M data bits to fill out the output code word. The output of the multiplexer 20 thus is (M+N) bits, wherein the N bits form the zero LSBs of the low gain signal path A and the highest MSBs of the high gain signal path B. In a typical application, the M bit A/D converters 10 and 12 are conventional 8 bit (M=8) flash-type A/D converters, and a gain of eight (i.e., N=3) is applied to the high gain path B, thereby resulting in an output of 11 (M+N) bits from the multiplexer 20. This results in two accuracy ranges; high accuracy at low signal levels and low accuracy at high signal levels (see FIG. 2). (The added accuracy N, is determined by the log base 2 of the gain in the amplifier prior to the lower A/D converter 12, while the base accuracy M is determined by the individual A/D converter. The resultant data word size is (M+N) bits.) This can be thought of as a crude approximation to the characteristics of the human visual system, and provides a near constant (quantization step size to input signal level) ratio after digitization.
Since the dual range A/D converter architecture requires signal dependent paths in order to perform as required, it is necessary that each signal path have similar electrical characteristics, be precisely aligned with respect to the other path, and have an A/D conversion crossover point that matches the theoretical transfer characteristic to within the system's required accuracy. For the dual range A/D converter shown in FIG. 1, the crossover point matching needs to be maintained to within half an LSB of M bits (plus or minus Vmax/((2**(M+1)-1))). The difficult task of calibrating the signal paths of a dual range A/D converter is complicated by the effects of temperature and long term stability of the signal path components.
The signal paths of the dual range A/D converter shown in FIG. 1 have transfer functions (Input Signal versus Output Digital Word for each individual A/D converter) as illustrated in FIG. 3. To maintain calibration of the dual range A/D converter, the transfer function of each converter path must have two points on the transfer curve precisely aligned with respect to similar points on the transfer curve of the other path. Assuming a linear transfer function, the first point will define the black level offset and the second point will set the slope or gain. In addition, the video signal levels in each path must be set in accordance with the A/D converter's operating range.
As can be seen from FIG. 3, the bottom end of the transfer curves requires that both signal paths of the dual range A/D converter output the same digital word (zero or some other desired digital offset above zero) when a black reference level is input from a sensor. This is accomplished by using either a digital or analog black level correction circuit, which automatically clamps on the reference black interval in the video (provided by dark pixels outside the image area in the case of a charge-coupled device (ccd) imager) and precisely sets the black reference level in each A/D converter path to the desired operating point. The analog black level correction method illustrated in FIG. 4 is disclosed in the earlier-cited paper by Oliphant and Weston. The two level-dependent signal paths A and B are followed by a dual sample and hold 30 and a single A/D converter 32 instead of using, as shown in FIG. 1, two signal paths, each having its own A/D converter. The proper path is selected by a comparator 34 driving a Flip-Flop 36 to generate a range indication signal that triggers the appropriate bit-shift after the A/D converter 32. The analog voltage of each path is sequentially sampled for the dark reference pixels in respective black level clamp feedback networks 38 and 40 and compared with a stable black reference voltage. A dc offset correction voltage is then generated, which is added to the video input (adders 42 and 44) to balance any inequality sensed by each comparator.
Black level correction can also be implemented digitally by comparing the digital data from the dark pixels in each A/D converter path with a black reference digital code, and either adding (positive or negative) a digital offset to the data word or feeding back an analog signal to offset the video input into the A/D converters. A known circuit for digital black level correction with analog feedback is illustrated in FIG. 5. A digital comparator 50, which is enabled during the black level correction period, compares the dark pixel code on its input A with a black reference digital code on its input B. If the dark pixel code is greater than the black reference digital code, then the output (A&gt;B) provides an offset voltage that is accumulated on a capacitor 52 and fed back to an subtractor 54 to decrease the level of the input signal. This brings the input black level down to the digital black reference. If the dark pixel code is less than the black reference digital code, then the output (A&lt;B) of the comparator 50 is inverted by an inverter 56 and applied to the capacitor 52, thereby discharging the capacitor 52 until the value subtracted from the video input brings the output of the subtracter 54 up to the black reference code. The dark pixels are typically available every line so that the capacitor 52 is brought to the black reference value for every line (or periodically for every few lines)
Black level correction satisfies the requirement for automatically establishing and matching the bottom end point of the transfer curves in each A/D converter path. In this sense, the known dual range A/D converter automatically self-calibrates for black level. However, it cannot be fully self-calibrating because it does not automatically match or stabilize a second point. In previous attempts to deal with this problem, the gain matching or setting of the second transfer curve point relied on high accuracy and low temperature coefficient components. As the requirements for digital word size M increased, these attempts have proven to be ineffective.
USPTO Backgrounds
USPTO Backgrounds