This application is a Continuation-in-Part of Application Ser. No. 10/460,622 Filed: May 24, 2002, Titled: DISTRIBUTED AVERAGING ANALOG TO DIGITAL CONVERTER TOPOLOGY, Inventors: MULDER et al. and is related to Application Ser. No. 10/158,774, Filed: May 31, 2002; Titled: ANALOG TO DIGITAL CONVERTER WITH INTERPOLATION OF REFERENCE LADDER, Inventors: MULDER et al. Application Ser. No. 10/158,193, Filed: May 31, 2002, Titled: CLASS AB DIGITAL TO ANALOG CONVERTER/LINE DRIVER, Inventors: Jan MULDER et al. and Application Ser. No. 10/158,733, Filed: May 31, 2002, Inventor: Jan MULDER; Titled: SUBRANGING ANALOG TO DIGITAL CONVERTER WITH MULTI-PHASE CLOCK TIMING, Inventors: van der GOES et al. all of which are incorporated by reference herein.
1. Field of the Invention
The present invention relates to analog to digital converters (ADC""s), and more particularly, to reducing nonlinearities and inter-symbol interference in high-speed analog to digital converters.
2. Related Art
A subranging analog to digital converter (ADC) architecture is suitable for implementing high-performance ADC""s (i.e. high speed, low power, low area, high resolution). FIG. 1 shows a generic two-step subranging architecture, comprising a reference ladder 104, a coarse ADC 102, a switching matrix 103, a fine ADC 105, coarse comparators 107, fine comparators 108 and an encoder 106. In most cases, a track-and-hold 101 is used in front of the ADC. In this architecture, an input voltage is first quantized by the coarse ADC 102. The coarse ADC 102 compares the input voltage against all the reference voltages, or against a subset of the reference voltages that is uniformly distributed across the whole range of reference voltages. Based on a coarse quantization, the switching matrix 103 connects the fine ADC 105 to a subset of the reference voltages (called a xe2x80x9csubrangexe2x80x9d) that is centered around the input signal voltage.
Modem flash, folding and subranging analog to digital converters (ADC""s) often use averaging techniques for reducing offset and noise of amplifiers used in the ADC. One aspect of averaging is the topology that is used to accomplish averaging, i.e., which amplifier outputs in which arrays of amplifiers are averaged together.
In general, flash, folding and subranging ADC""s use cascades of distributed amplifiers to amplify the residue signals before they are applied to the comparators. These residue signals are obtained by subtracting different DC reference voltages from an input signal Vin. The DC reference voltages are generated by the resistive ladder (reference ladder) 104 biased at a certain DC current.
High-resolution ADC""s often use auto-zero techniques, also called offset compensation techniques, to suppress amplifier offset voltages. In general, autozeroing requires two clock phases (xcfx861 and xcfx862). During the auto-zero phase, the amplifier offset is stored on one or more capacitors, and during the amplify phase, the amplifier is used for the actual signal amplification.
Two different auto-zero techniques can be distinguished, which are illustrated in FIGS. 2 and 3. The technique shown in FIG. 2 connects an amplifier 201 in a unity feedback mode during the auto-zero clock phase xcfx861. As a result, a large part of the amplifier 201 input offset voltage is stored on input capacitors C1a, C1b. The remaining offset is stored on output capacitors C2a, C2b if available.
The second technique, shown in FIG. 3, shorts the amplifier 201 inputs during the auto-zero phase xcfx861 and connects them to a DC bias voltage Vres. Here, the amplifier 201 output offset voltage is stored on the output capacitors C2a, C2b. Many ADC architectures use a cascade of several (auto-zero) amplifiers to amplify the input signal prior to applying to the comparators 107, 108. In general, flash, folding and subranging ADC""s use arrays of cascaded amplifiers, and averaging and interpolation techniques are used to improve performance.
Unfortunately, the performance of cascaded arrays of amplifiers degrades significantly at high clock and input signal frequencies. The cause of this degradation is illustrated in FIG. 4 when the reset technique shown in FIG. 3 is used, and where RSW is shown as a circuit element, and the current flow IC is explicitly shown.
When the amplifier 201 is in the auto-zero phase xcfx861, the input capacitors C1a, C1b are charged to the voltage Vsample that is provided by the track-and-hold amplifier 101. As a result, a current IC will flow through the input capacitors C1a, C1b and an input switch (not shown). Due to the finite on-resistance RSW of the input switch (see FIG. 4), an input voltage is generated, which will settle exponentially towards zero. This input voltage is amplified by the amplifier 201 and results in an output voltage that also slowly settles towards zero (assuming the amplifier 201 has zero offset).
Essentially, the auto-zero amplifier 201 is in a xe2x80x9cresetxe2x80x9d mode one-half the time, and in an xe2x80x9camplifyxe2x80x9d mode the other one-half the time. When in reset mode, the capacitors C1a, C1b are charged to the track-and-hold 101 voltage, and the current IC flows through the capacitors C1a, C1b and the reset switches, so as to charge the capacitors C1a, C1b. 
When the ADC has to run at high sampling rates, there is not enough time for the amplifier 201 output voltage to settle completely to zero during the reset phase. As a result, an error voltage is sampled at the output capacitors C2a, C2b that is dependent on the voltage Vsample. This translates into non-linearity of the ADC, and often causes inter-symbol interference (ISI).
The problem of ISI occurs in most, if not all, ADC architectures and various approaches exist for attacking the problem. The most straightforward approach is to decrease the settling time constants. However, the resulting increase in power consumption is a major disadvantage.
Another approach is to increase the time allowed for settling, by using interleaved ADC architectures. However, this increases required layout area. Furthermore, mismatches between the interleaved channels cause spurious tones. The ISI errors can also be decreased by resetting all cascaded amplifiers during the same clock phase. Unfortunately, this is not optimal for high speed operation either.
The present invention is directed to an analog to digital converter topology that substantially obviates one or more of the problems and disadvantages of the related art.
There is provided an analog to digital converter including a reference ladder, a clock having phases xcfx861 and xcfx862, and a track-and-hold amplifier tracking an input signal with its output signal during the phase xcfx861 and holding a sampled value during the phase xcfx862. A plurality of coarse amplifiers each input a corresponding tap from the reference ladder and the output sign. A plurality of fine amplifiers input corresponding taps from the reference ladder and a signal corresponding to the output signal, the taps selected based on outputs of the coarse amplifiers. A circuit responsive to the clock receives the signal corresponding to the output signal, the circuit substantially passing the signal corresponding to the output signal and the corresponding taps to the fine amplifiers during the phase xcfx862 and substantially rejecting the signal corresponding to the output signal during the phase xcfx862. An encoder converts outputs of the coarse and fine amplifiers to an N-bit digital signal representing the input signal.
In another aspect of the present invention there is provided an analog to digital converter including a reference ladder and a two-phase clock having phases xcfx861 and xcfx862. A track-and-hold amplifier tracking an input signal with its output signal during the phase xcfx861 and holding a sampled value during the phase (xcfx862. A plurality of coarse amplifiers each inputting a signal corresponding to the output signal and a corresponding tap from the reference ladder. A switching circuit that receives the signal corresponding to the output signal and has a differential mode transfer function of approximately 1 on the phase xcfx862 and approximately 0 on the phase xcfx861. A plurality of fine amplifiers inputting corresponding taps from the reference ladder and the signal corresponding to the output signal through the switching circuit, the taps selected based on outputs of the coarse amplifiers. An encoder converts outputs of the coarse and fine amplifiers to an N-bit digital signal representing the input signal.
In another aspect of the present invention there is provided an analog to digital converter including a reference ladder and a multi-phase clock. A track-and-hold amplifier tracking an input signal with its output signal during one phase of the multi-phase clock and holding a sampled value during another phase of the multi-phase clock. A plurality of coarse amplifiers each inputting a signal corresponding to the output signal and a corresponding tap from the reference ladder. Switching means that receives the signal corresponding to the output signal and responsive to the multi-phase clock, the means substantially passing the signal corresponding to the output signal to the fine amplifiers during the one phase and substantially rejecting the signal corresponding to the output signal during the another phase. A plurality of fine amplifiers inputting, through the switching means, corresponding taps from the reference ladder and the output signal, the taps selected based on outputs of the coarse amplifiers. An encoder converts outputs of the coarse and fine amplifiers to an N-bit digital signal representing the input signal.
An analog to digital converter including a track-and-hold amplifier whose output signal tracks an input signal during one clock phase, and holds a sampled value during another clock phase. A coarse amplifier inputting the output signal and a coarse tap. A transfer matrix that substantially passes a signal corresponding to the output signal during the one clock phase and substantially blocks the signal corresponding to the output signal during the another clock phase. A fine amplifier inputting a fine tap and the output signal through the transfer matrix, the fine tap selected based on an output of the coarse amplifier. An encoder converts outputs of the coarse and fine amplifiers to an N-bit digital signal representing the input signal.
An analog to digital converter including a differential coarse amplifier inputting a signal corresponding to an input signal and a coarse tap during one clock phase, and a sampled value during another clock phase. A plurality of cross-coupled transistors that substantially pass the first signal and a fine tap during the one clock phase and substantially block the first signal and the fine tap during the another clock phase, the fine tap selected based on a signal from the differential coarse amplifier. A differential fine amplifier inputting an output of the plurality of cross-coupled transistors. An encoder converts outputs of the coarse and fine amplifiers to an N-bit digital signal representing the input signal.
Additional features and advantages of the invention will be set forth in the description which follows, and in part will be apparent from the description, or may be learned by practice of the invention. The advantages of the invention will be realized and attained by the structure particularly pointed out in the written description and claims hereof as well as the appended drawings.
It is to be understood that both the foregoing general description and the following detailed description are exemplary and explanatory and are intended to provide further explanation of the invention as claimed.
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