1. Field of the Invention
The present invention relates to switched power supplies, and more specifically, to a circuit for a boost converter which incorporates a switched shunt inductance. The shunt inductance is used to produce a zero volt switching condition across the primary power switch of the power supply. This permits the primary switch to be resonantly switched on with a zero voltage condition in conjunction with a large value for the resonant snubber capacitor. These features reduce the power losses associated with changing the state of the primary switch compared to conventional zero volt switching circuits, and improve the overall efficiency of the converter. The circuit techniques can be applied to continuous and discontinuous boost high power, and high and constant frequency regulators.
2. Description of the Prior Art
Switching or "switch mode" power supplies use a semiconductor device as a power switch to control the application of a voltage to a load. A boost or step-up converter is used to produce an output (or load) voltage which is of the same polarity but a higher voltage than the input voltage (V.sub.in) supplied by the input power supply (not shown). FIG. 1 is a schematic diagram showing a basic circuit for a prior art boost converter 100. The operation of power switch Q2 102 is controlled by applying a waveform to gate node 103. When switch Q2 is turned "on", i.e., conducting, the input voltage V.sub.in is applied across primary inductor L 104. Under steady-state conditions, the current (I) in inductor L will increase linearly with time to a peak value as energy from the input supply is stored in its magnetic core, as described by the relationship V=L dI/dt. Rectifier D1 106 will be reversed biased and thus not conducting. In addition, under steady-state conditions, current will flow from output capacitor C.sub.0 108 to the load attached across the output terminals of boost converter 100 as a function of the output voltage (V.sub.out). This causes the output capacitor to discharge.
When switch Q2 is turned "off", i.e., not conducting, current is no longer supplied by the input supply to inductor L. The inductor attempts to compensate for this change by causing the voltage across the inductor to increase, raising the potential at node A. This keeps the current in L flowing in the same direction as before. When the voltage at node A exceeds the sum of the output voltage across C.sub.0 and the diode drop across diode D2, rectifier D2 becomes forward biased and hence conducting. This transfers the current in inductor L to output capacitor C.sub.0 and the load. Since the output voltage is greater than the supply voltage, inductor L becomes reverse-biased and the current in it decays linearly downward to its original value. Power switch Q2 is then turned back on to start another cycle. Just before being turned on, the voltage across the power switch is greater than the input voltage, V.sub.in. As the power switch is turned on, the voltage falls and current through the switch increases, resulting in a loss of power. The output voltage, V.sub.out, is determined by the duty ratio of power switch Q2 and the supply voltage, V.sub.in, according to the following formula: EQU V.sub.out =V.sub.in /(1-D),
where D is the duty ratio of the switch and is defined as t.sub.on /(t.sub.on +t.sub.off), with t.sub.on being the "on" time of the switch during a cycle and t.sub.off being the "off" time during a cycle.
A drawback of switch mode power circuits as above described is that the switching devices in such switch mode power converters are subjected to high stresses and potentially high switching power loss as a result of the switch being changed from one state to another while having a significant voltage across it. These effects increase linearly with the switching frequency of the waveform used to control the power switch. Another drawback of switched power circuits is the electromagnetic interference arising from the large dI/dt and dV/dt that occurs when the switch changes state.
The noted disadvantages of switch mode power converters can be reduced if each power switch in the circuit is caused to change its state (from "on" to "off" or vice versa) when the voltage and/or current through it is zero or at a minimum. Such a control scheme is termed "zero-voltage" and/or "zero-current" switching. In the case of switching at a minimum voltage, the control scheme is termed "low-voltage" switching. It is thus desirable to switch the power switching device(s) at instances of zero or minimum voltage in order to reduce stress on the switch(es) and power loss of the power supply or converter. This increases the efficiency of the power supply or converter.
One method of implementing zero voltage switching is to provide a voltage signal across the power switch which passes through a zero value. This can be done by connecting a resonant network (typically an inductor and a capacitor) to the power supply circuit. The network acts to smooth the output signal of the power supply and provide a back emf across the power switch in the form of a sinusoidally varying waveform. The resonant elements are arranged so that the back emf waveform generates a zero crossing voltage signal across the power switch while the switch is off. This provides a zero-voltage or zero-current condition through the power switch which can be used to define the desirable switching point(s) for turning the switch on.
FIG. 2 is a schematic drawing of the boost converter of FIG. 1 to which has been added an LC resonant network to provide a zero volt switching condition. As is shown in the figure, a resonant capacitor C.sub.res 110 is connected across power switch Q2 and a resonant inductor L.sub.res 112 is connected between C.sub.res and rectifier D2. When switch Q2 is on, the operation of the circuit of FIG. 2 is the same as that described for the boost converter of FIG. 1. No current flows through L.sub.res because rectifier D2 is not conducting.
When switch Q2 is off, current is transferred from primary inductor L to C.sub.res, charging the capacitor. After C.sub.res becomes charged, the potential at node A increases until it exceeds the output voltage across C.sub.0. At this point, rectifier D2 becomes forward biased and hence conducting. This transfers current from inductor L to L.sub.res and into output capacitor C.sub.0 and the load. As the current in L declines, the current being forced into L.sub.res also declines. L.sub.res responds by pulling charge from capacitor C.sub.res, L.sub.res and C.sub.res form a resonant circuit which produces a sinusoidally changing voltage signal across switch Q2, providing a zero crossing signal which can be used to define the desired switching point.
FIG. 3 is a schematic drawing of a boost converter to which has been added an LC resonant network which is shunted to a ground reference by the action of a second switch Q1 118. In FIG. 3, inductor L.sub.res 112, is connected to ground via an amorphous core 111, an active switch Q1, and diode 114. There is a shunt path around switch Q1 to ground formed by a resistor 117 and a diode 115.
Amorphous core 111 is typically required, with a ground reference for switch Q1 because after the energy in inductor 112 is put back into output capacitor 108 via diode 116, the current in inductor 112 will turn off and can swing the voltage on the junction between inductor 112 and the cathode of diode 115 to ground, causing diode 115 and resistor 117 to conduct continuously. Resistor 117 allows any residual energy in L.sub.res to dissipate quickly.
FIGS. 4(A) to 4(F) are timing diagrams illustrating the operation of the boost converter of FIG. 3. FIG. 4(A) shows the current in inductor L.sub.res 112, indicating that the current builds up to the level of that in diode 106 (I.sub.D2) which has been conducting during the period t.sub.0 to t.sub.1 after the grounded shunt switch Q1 is turned on at t.sub.0, as shown in FIG. 4(E), which illustrates the gate drive signal for Q1. After time t.sub.1 the current continues to build up in inductor 112 as the voltage on switch Q2, (shown in FIG. 4(C)) and the voltage at node 119 change in a resonant manner to discharge capacitor C.sub.res 110 during the period between t.sub.1 and t.sub.2.
At time t.sub.2, diode 101, which is connected in parallel with switch Q2 102, conducts, as the energy in L.sub.res is clamped since there can be no further voltage change on C.sub.res. The level of this reverse current corresponds to the build up of current above the main load current in inductor 104, which occurs because, during the period t.sub.1 to t.sub.2, the voltage across L.sub.res 112 is positive and therefore stores extra energy during this period. Initially all the current in L.sub.res 112 will conduct in body diode 101, but this changes as the current from main inductor L 104 equalizes part of the current. The main drive signal to switch Q2 102 (shown in FIG. 4(F)) will start soon after t.sub.2, with a propagation delay of approximately 50 to 70 nanoseconds. The gate drive to switch Q1, (shown in FIG. 4(E)) is terminated at t.sub.3. Switch Q1 is required to switch a large current (see V.sub.-- Turnoff in FIG. 4(B) and I.sub.-- Turnoff in FIG. 4(A)), which creates both a noisy and lossy situation. The turnoff for shunt switch Q1 cannot be extended because the voltage across inductor L.sub.res 112 during the period between t.sub.2 and t.sub.3 is very small since both Q1 and Q2 are clamped to Ground, causing the reduction in current, -DI/DT (shown in FIG. 4(A)), to be very slow. If the shunt switch Q1 was not turned off but instead kept on, the current would staircase up in L.sub.res until saturation occurred.
The amount of excess energy stored in inductor L.sub.res 112, is sent to output capacitance C.sub.0, 108, via diode 116, as illustrated by the hatched area of FIG. 4(A). FIGS. 4(A) and 4(B) further illustrate the current decrease in L.sub.res (4(A)) and the recovery of the voltage on switch Q1 (4(B)), between t.sub.3 and t.sub.4. As can be seen from FIG. 4(B), the voltage on switch Q1 can reverse, and amorphous core 111 is used to prevent this from occurring.
In Zero Voltage Switching (ZVS) power converters, during each switching cycle the voltage across the power switch is driven to zero by the action of the inductive load, and ideally, the switch is then turned on. This typically requires that ZVS Resonant converters have a large LC tank to ensure that there is sufficient inductive energy to drive the voltage across the switch to zero. However, a disadvantage of this means of providing a zero voltage signal across the power switch is that there is significant power loss associated with the large intrinsic resistance of the resonant network capacitance and inductance, with the power loss being approximately proportional to the values of those elements. This reduces the efficiency of the converter.
What is desired is a boost converter circuit which is capable of zero voltage power switch operation, where the circuit elements responsible for the zero voltage switching produce a lower power loss than presently available devices.
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