1. Field of the Invention
The present invention relates to a frequency changer. Such a frequency changer may, for example, be used as part of a zero intermediate frequency (ZIF) tuner, for instance, for a digital direct broadcasting by satellite (DBS) receiver system. The present invention also relates to a digital tuner. In general, the invention may be applied to any tuner where a received channel contains a local oscillator frequency.
2. Description of the Prior Art
A known type of digital tuner for use in a DBS system is illustrated in FIG. 1 of the accompanying drawings and is based on the well-known super-heterodyning technique, which has been used in analogue tuners for many decades. An antennae input 1 for receiving an input signal from an antennae system is connected to the input of a radio frequency (RF) amplifier 2, which comprises a low noise amplifier 3 and an automatic gain control (AGC) arrangement 4. The RF amplifier 2 is connected to a frequency changer via a tracking bandpass filter 60 tuned to pass the selected channel.
The RF amplifier 2 and the filter 60 form a “front end” which is external to the frequency changer for down-converting the input RF signal to a lower fixed intermediate frequency (IF) which is typically 402 or 480 MHz. The frequency changer comprises a first integrated circuit 5 which comprises an amplifier 6, a multiplier 7, an amplifier 8, and an oscillator 9. The oscillator 9 has an off-chip variable tuned circuit 10 which determines the frequency of oscillation of the oscillator 9 and whose resonant frequency is controlled by a phase-locked loop (PLL) frequency synthesiser 11. The synthesiser 11 also controls tuning of the filter 60, which is offset from the frequency of the oscillator by the IF.
The output of the frequency changer is connected via a bandpass surface acoustic wave (SAW) filter 12 to a quadrature down-converter 13. The down-converter 13 comprises an amplifier 14 which supplies the filtered IF signal to multipliers 15 and 16 whose outputs are connected via amplifiers 17 and 18, respectively, to in-phase (I) and quadrature (Q) outputs. A reference oscillator 19 generates a signal whose frequency is equal to the intermediate frequency. This signal is supplied to the multipliers 15 and 16 via a phase adjusting network 20 so that the multipliers 15 and 16 receive signals which are in quadrature i.e. 90° out of phase which each other.
The oscillator 19 has a frequency-determining tuned circuit 21 which is connected to receive a Costas feedback signal from a Costas de-rotation circuit in a quadrature phase shift keyed (QPSK) demodulator to which the I and Q outputs of the down-converter 13 are connected. Such a demodulator with Costas de-rotation circuitry is well known.
In use, the input signal is amplified and gain-controlled by the RF amplifier 2 and converted by the frequency changer to the IF signal. The synthesiser 11 provides tuning for selecting the frequency of the input signal to be received and demodulated. The down-converter 13 converts the IF signal to baseband in-phase and quadrature output signals which are supplied to the demodulator (not shown). The demodulator supplies the Costas feedback signal to the tuned circuit 21 of the reference oscillator 19 so as to control the phase of the output signal of the oscillator 19 and hence the phases of the quadrature signals supplied to the multipliers 15 and 16. This control loop ensures that the I output supplies the in-phase signal substantially uncontaminated with the quadrature signal whereas the Q output supplies the quadrature signal substantially uncontaminated by the in-phase signal.
Although this known type of digital tuner provides acceptable performance, it is relatively complicated and relatively expensive to manufacture. For example, the parts of the timer illustrated in FIG. 1 have to be fabricated with several integrated circuits and with an external front end and often require alignment of the tracking filter.
Another known type of digital tuner for use in DBS receivers differs from that illustrated in FIG. 1 in that the de-rotation function is performed by subsequent digital signal processing (DSP) techniques. The Costas feedback signal to the tuned circuit 21 is thus unnecessary and the reference oscillator 19 is free-running i.e. not phase-locked. The output signals from the amplifiers 17 and 18 each therefore contain both in-phase and quadrature signals which are subsequently extracted by digital signal processing.
FIG. 2 of the accompanying drawings illustrates another known type of digital tuner for use in a DBS receiving system. The tuner is of the ZIF type and converts the input signal in a single conversion step to the base-band signals. The tuner illustrated in FIG. 2 differs from that illustrated in FIG. 1 in that the frequency changer and down-converter are combined into a single frequency changer stage 13, which comprises the amplifiers 14, 17, 18 and multipliers 15 and 16 with the oscillator 9, the variable tuned circuit 10 and the synthesiser 11. The bandpass filter 12 is eliminated and the I and Q outputs are supplied via low-pass filters 22 and 23.
The frequency of the oscillator 9 set by the synthesiser 11 is at or very near the centre frequency of the channel containing the desired input signal as supplied by the RF amplifier 2. Local oscillator signals are supplied in quadrature by the phase adjusting network 20 to the first and second multipliers 15 and 16, which supply base-band demodulated signals via the low-pass filters 22 and 23 to subsequent digital signal processing circuitry for performing de-rotation to retrieve the orthogonal in-phase and quadrature modulation signals.
Although the ZIF tuner shown in FIG. 2 represents a substantial simplification compared with the tuner shown in FIG. 1, it is still not possible to form all of the main tuner circuitry in a single integrated circuit and achieve the necessary performance for acceptable results.
The problems with such a tuner architecture result from the fact that the local oscillator frequency in the frequency changer 13 is within the frequency band or channel of the desired input signal. Because the two signals are of substantially the same frequency, any signal leakage which inevitably occurs results in interference between the signals as explained below.
Leakage of the local oscillator signal to the RF input has two effects. First, such leakage results in re-radiation of the local oscillator signal to other tuners which may be connected to the same antenna system or may be located nearby. Although this problem may be substantially overcome by providing an external front end as illustrated at 2 in FIG. 2 having sufficient “reverse isolation” to meet tuner re-radiation specification requirements, it is then impossible to provide an “internal” RF amplifier 2, for example within the integrated circuit in which the main parts of the frequency changer 13 are formed.
Second, local oscillator signals leaking to the input are amplified and supplied with the input signal to the frequency changer 13. Because of phase shifts which are inherent in such leakage mechanisms and because of amplification within the ZIF tuner, this results in a DC imbalance at the outputs of the frequency changer 13. Such a DC imbalance can represent a significant fraction of the desired signal and cannot be improved because of limits to the degree of isolation which is achievable.
Leakage also occurs from the RF input to the local oscillator and this results in “injection pulling” of the local oscillator 9. This in turn degrades the phase noise and hence the quality of the signal supplied by the oscillator 9 to the multipliers 15 and 16, resulting in reduced signal-to-noise performance of the tuner. Injection pulling results from the injection of the RF signal into the oscillator, which is in effect a tuned amplifier with a low Q feedback network at the oscillation frequency. The injected RF signal is amplified within the oscillator loop and effectively impresses its characteristic on the output of the reference oscillator 9. The RF signal carries a pseudo random noise (PSRN) type modulation and appears like broadband noise, which is impressed on the oscillator signal and thus degrades the noise performance.
The degradation caused by injection pulling to the oscillator phase noise performance thus limits the application of ZIF techniques within DBS applications because lower data rates require a very pure local oscillator signal in order to achieve satisfactory performance. In practice, in order to achieve satisfactory performance with ZIF techniques, a relatively low amplitude input signal must be supplied to the input of the ZIF circuit and this results in disadvantages such as reduced dynamic range. Alternatively or additionally, complex techniques must be used in order to suppress injection pulling so as to achieve the desired standard of performance.
It is known in other applications to form the reference oscillator “on chip” so as to reduce oscillator coupling, for which the dominant leakage mechanism is through parasitic components in integrated circuit packaging and by electromagnetic coupling to and from oscillator strip lines. Such techniques have been used, for example, in global positioning satellite (GPS) receiver systems, for example as disclosed in Schaeffer Shahani et al, “A 115 mW CMOS GPS Receiver”, ISSCC98 paper Ref. FA 8.1. However, such integrated oscillators have such inherently poor phase noise characteristics that they are totally unsuitable for use in DES receiving systems. Also, such techniques have been applied to conventional super-heterodyning architectures but have not been applied to ZIF architectures.
It is well-known that an oscillator with poor phase noise performance can be improved by applying a phase-locked loop to control the phase noise within the loop bandwidth. Such techniques are, for example, disclosed in “Digital PLL Frequency Synthesisers”, Rohde, Prentice Hall, 1983, ISBN 013 214 293-2. However, this provides no improvement in phase noise outside the loop bandwidth. The maximum loop bandwidth which may be applied is determined by the required reference oscillator frequency step size and is of the order of a few kilohertz. Thus, the integrated phase noise outside this bandwidth has a significant detrimental effect on performance.
GB 2 319 913 discloses a superheterodyne receiver of the double conversion type. A local oscillator for the first mixer is phase-locked via a fixed frequency divider to a local oscillator for the second mixer. The second oscillator is, in turn, phase-locked to a reference frequency.
U.S. Pat. No. 4,607,393 discloses a receiver for receiving and demodulating multiplexed stereo signals which art frequency modulated on a carrier. The receiver is of the single conversion type which converts the incoming broadcast band frequencies to an intermediate frequency of 10.7 MHz. The mixer receives a local oscillator signal which is phase-locked to a further local oscillator. The further local oscillator is phase-locked at a multiple of the frequency of the pilot tone of the demodulated stereo multiplexed signal.
EP 0 253 680 discloses a receiver for receiving and demodulating an angle modulated signal of a mobile subscriber set. The receiver is of the double conversion type with a first mixer receiving the output of a local oscillator which is phase-locked to a temperature controlled crystal oscillator. The crystal oscillator is capable of being tuned over a narrow frequency range, which is controlled with reference to the received signal after the second conversion but prior to demodulation.
USPTO Backgrounds
USPTO Backgrounds