(1) Field of the Invention
The present invention relates to semiconductor devices which are used and embedded in spread spectrum radar apparatuses employing a spread spectrum scheme, and more particularly to a semiconductor device which is used in such spread spectrum radar apparatuses and suppresses spurious signals caused by non-linear components of active elements.
(2) Description of the Related Art
In recent years, radar apparatuses have been equipped in vehicles to detect preceding vehicles, obstacles located backward, and the like. This provides significant achievements, including safety improvement such as collision avoidance, enhancement of driving convenience represented by reverse driving support, and the like. In accordance with the achievements, various technologies regarding radar apparatuses equipped in vehicles (hereinafter, referred to as “in-vehicle radar apparatuses”) have been proposed. One example of such technologies is a radar apparatus using a spread spectrum scheme (hereinafter, referred to as a “spread spectrum radar apparatus”) (see Japanese Unexamined Patent Application Publication No. 7-12930, for example).
For such an in-vehicle radar apparatus, it is critical to suppress influence of radio waves transmitted from same- or similar-type radar apparatuses equipped in other vehicles. In order to achieve this, the spread spectrum radar apparatus transmits radio waves which are received and suppressed by different-type radar apparatuses using different code sequences or different schemes. Thereby, the spread spectrum radar apparatuses do not much affect radar apparatuses of different types. Moreover, the spread spectrum radar apparatuses do not have any serious troubles for their object detection ability, even if undesired radio waves are transmitted from other spread spectrum radar apparatuses or radio communication apparatuses using the same frequency band.
This is because, in the spread spectrum radar apparatuses, the radio waves whose frequency is spectrum-spread over a wider band using pseudo noise codes (hereinafter, referred to as “PN codes”) are transmitted. Further, since radio waves are spectrum-spread over a wider band, power consumption per unit frequency is reduced and the influence to other radar apparatuses is decreased. Furthermore, by adjusting a chip rate and a code period of the PN code, a relationship between distance resolution and the maximum detectable distance is set flexibly and the radio waves are thereby transmitted continuously, so that peak power is not increased.
FIG. 1 is a diagram showing a structure of the conventional spread spectrum radar apparatus. The conventional spread spectrum radar apparatus 10 shown in FIG. 1 includes a clock signal generation unit 11, a PN code generation unit 12, a code delay unit 13, a signal source 21, a spread spectrum modulation unit 22, a transmitting unit 23, a transmission antenna 24, a receiving antenna 31, a receiving unit 32, an inverse spread spectrum modulation unit 33, and a signal processing unit 34. Here, it is assumed that the spread spectrum radar apparatus 10 uses a M-sequence code as the PN code. As shown in FIG. 1, it is assumed that the conventional spread spectrum radar apparatus 10 uses autocorrelation characteristics that a M-sequence code has a single peak. Using the autocorrelation characteristics, a delay time period is varied to detect the delay time period during which the single peak is obtained, thereby measuring reflection intensity at an object and a distance to the object.
In the spread spectrum radar apparatus 10, the signal source 21 generates a narrow-band signal, and the spread spectrum modulation unit 22 performs spread-spectrum modulation on the narrow-band signal using a PN code generated by the PN code generation unit 12, in order to generate a broad-band signal. Then, the spread spectrum modulation unit 22 outputs the resulting broad-band signal to the transmission antenna 24 via the transmitting unit 23. The transmission antenna 24 transmits the obtained broad-band signal as detection radio waves.
Moreover, in the spread spectrum radar apparatus 10, the receiving antenna 31 receives reflected waves of the detection radio waves which have been transmitted and then reflected at an object. The receiving antenna 31 provides, as received signal, the reflected waves to the inverse spread spectrum modulation unit 33 via the receiving unit 32. The inverse spread spectrum modulation unit 33 performs inverse spread-spectrum modulation on the received signal, using a PN code which is delayed by the code delay unit 13. Then, the inverse spread spectrum modulation unit 33 provides the resulting signal to the signal processing unit 34.
Here, it is examined the case where the signal provided by the inverse spread spectrum modulation unit 33 has the same frequency components as the narrow-band signal generated by the signal source 21. In this case, the signal processing unit 34 determines the time period which is delayed by the code delay unit 13, as a time period during which the detection radio waves have been transmitted and returned (hereinafter, referred to as a “reciprocating propagation time period”). In addition, a distance corresponding to the determined reciprocating propagation time period is determined as a distance to the object.
On the other hand, it is examined the case where the signal provided by the inverse spread spectrum modulation unit 33 does not have the same frequency components as the narrow-band signal generated by the signal source 21. In the case, the signal processing unit 34 changes a time period delayed by the code delay unit 13.
Here, the inverse spread spectrum modulation unit 33 generally includes a Binary Phase Shift Keying modulator (BPSK modulator) such as a balanced mixer.
FIG. 2 is a diagram showing a circuit configuration of the inverse spread spectrum modulation unit (semiconductor device) of the conventional spread spectrum radar apparatus. As shown in FIG. 2, the inverse spread spectrum modulation unit 33 of the conventional spread spectrum radar apparatus includes a balanced inverse spread spectrum circuit 61 and an unbalanced to balanced transforming circuit 62. The unbalanced to balanced transforming circuit 62 includes a current power circuit which supplies bias currents to operate transistors included in the balanced inverse spread spectrum circuit 61. The balanced inverse spread spectrum circuit 61 includes transistors Q1, Q2, Q3, and Q4, resistors R1 and R2, a power source Vc, output terminals OUT1 and OUT2, and pseudo-noise (PN) code terminals PN1 and PN2. The unbalanced to balanced transforming circuit 62 includes transistors Q5, Q6, and Q7, capacitors C1 and C2, resistors R3, R4, and R5, a received signal terminal RF, and power sources Vb1 and Vb2. The balanced inverse spread spectrum circuit 61 and the unbalanced to balanced transforming circuit 62 are connected with each other via a balanced line 51 including a line 51a and a line 51b. The inverse spread spectrum modulation unit 33 is well-known as a Gilbert cell. The inverse spread spectrum modulation unit 33 is designed so that delay does not occur between a pair of a balanced signal consisting of a positive signal and a negative signal (hereinafter, referred to also as “balanced signal pair).
In more detail, in the inverse spread spectrum modulation unit 33, a base of the transistor Q5 is connected to the received signal terminal RF via the capacitor C1, and a base of the transistor Q6 is connected to ground at high frequency via the capacitor C2. A collector of the transistor Q5 is connected the line 51a, and a collector of the transistor Q6 is connected to the line 51b. When an unbalanced signal is inputted from the received signal terminal RF, the inputted unbalanced signal is converted into a balanced signal pair which is then outputted to the balanced line 51.
Furthermore, in the inverse spread spectrum modulation unit 33, bases of the transistors Q1 and Q4 are connected to the PN code terminal PN1, and bases of the transistors Q2 and Q3 are connected to the PN code terminal PN2. Collectors of the transistors Q1 and Q3 are connected to the output terminal OUT1, and collectors of the transistors Q2 and Q4 are connected to the output terminal OUT2. Emitters of the transistors Q1 and Q2 are connected to the line 51a, and emitters of transistors Q3 and Q4 are connected to the line 51b. When the balanced signal pair is inputted via the balanced line 51 to a switch circuit which includes the transistors Q1, Q2, Q3, and Q4, polarities of the inputted balanced signal pair is reversed depending on the PN codes which are inputted as a differential signal from the PN code terminals PN1 and PN2. The positive and negative signals in the pair whose polarities are reversed are outputted from the output terminals OUT1 and OUT2, respectively.
Unfortunately, following problem is encountered in the conventional technology. Here, it is assumed that the detection radio waves have been transmitted for a predetermined time period and then the transmitted detection radio waves have been reflected at a plurality of objects, so that the receiving antenna 31 receives plural kinds of reflected waves having respective different reciprocating propagation time periods. Then, a plurality of different received signals are inputted into the inverse spread spectrum modulation unit 33, and thereby the signals outputted from the inverse spread spectrum modulation unit 33 have distortion, which results in a problem that spurious signals occur where signals do not exist originally.
This problem is caused by the following reasons. Regarding the transistor Q5 to which the plurality of received signals are inputted, a collector current (IC) is exponentially varied depending on a collector-to-emitter voltage (VCE), as obvious from the characteristics (IC-VCE characteristics) between the collector current (IC) and the collector-to-emitter voltage (VCE). Thereby, the signals outputted from the transistor Q5 include high-order components of the inputted signals, and these high-order components occur as the distortion.
More specifically, the high-order components cause components generated by multiplication operations among the plurality of the received signals. By shift additivity of M-sequence codes, the generated components become a signal that is equivalent to the signal which is generated by spread-spectrum modulation using the same M-sequence code and which has the third delay amount different from the delay amount of each of the received signals. Thereby, in the signals outputted from the inverse spread spectrum modulation unit 33, a spurious signal occurs as if a reflecting object is located at a position which corresponds to the third delay amount and at which such an object is not located actually. As a result, inconvenience occurs.
Note that the shift additivity of M-sequence codes means characteristics that, when an EXCLUSIVE-OR operation is performed on two M-sequence codes which have different delay amounts but are generated by the same M-sequence code generator, a M-sequence code is generated which has the third delay amount different from the delay amounts of the former M-sequence codes.
Furthermore, the conventional technology encounters another problem. Since only a base of the transistor Q6 is connected to ground at high frequency, the transistors Q5 and Q6 are operated in an unbalanced state. In such a state, the balanced signal pair outputted from the transistors Q5 and Q6 has even-order components which are originally suppressed in a differential circuit. Thereby, undesired signals resulted from the shift additivity of M-sequence codes are generated, so that the signals outputted from the inverse spread spectrum modulation unit 33 include spurious signals.
USPTO Backgrounds
USPTO Backgrounds