All digital, narrowband radio transmitters that are spectrally efficient require, in principle, two operations to be performed: (1) the baseband data must be filtered to limit the width of its spectrum, and (2) the resulting baseband signal must be translated to the desired radio frequency band. A number of techniques exists for translating the baseband signal to the radio frequency signal. One technique involves feeding the baseband signal directly into the inputs of a frequency synthesizer, such as a PLL (phase-locked loop).
Operation of the PLL is well known to persons having ordinary skill in this field and will therefore not be described here. It will suffice to say that the division factor N of the PLL can be either an integer value or it can be a non-integer value, i.e., a fractional-N PLL. Fractional-N PLLs are usually controlled by sigma delta modulators. The sigma delta modulator switches the division factor of the PLL between different integer values such that the resulting average value of the PLL output signal can be made a fractional multiple of its reference signal.
Applying a baseband signal to the sigma delta modulator results in direct modulation of the fractional-N PLL. Typically, a filtered version of the baseband signal is provided to the sigma delta modulator, which then uses the instantaneous frequency of the baseband signal to vary the frequency division factor of the frequency divider. By controlling the frequency division factor with a sigma-delta modulator, modulation with a constant envelope (i.e., frequency and phase modulation) can be generated. And because the sigma delta modulator takes the place of complicated analog circuitry, extremely compact architectures can be developed for constant envelope systems (e.g., Global System for Mobile Communications (GSM) or Digital Communication Systems (DCS)). Currently, a complete radio transmitter may be integrated into a single ASIC (application specific integrated circuit) using the direct modulation approach.
Constant envelope systems are not bandwidth efficient, however, and therefore some proposed systems also use amplitude modulation in addition to phase and frequency modulation. Examples of these systems include EDGE (Enhanced Data GSM Environment) and WCDMA (Wideband Code Division Multiple Access). In these systems, the modulating signal is divided into a phase part and an amplitude part. The phase part is introduced in the fractional-N PLL and the amplitude part is added (effectively multiplied) in a post PLL power amplifier. In this way, switching blocks can be used throughout the complete modulator, which is very power efficient.
When dividing the signal into an amplitude and a phase part, however, the respective bandwidth of the phase and of the amplitude part become much wider than that of the combined signal. And since the amplitude and the phase part are combined in a multiplier after the PLL, stringent requirements are imposed on the dynamic range and bandwidth of the amplitude and phase parts, and also on the timing between the amplitude and phase parts.
One way to get around the PLL loop bandwidth limitation is to add another modulation point to the PLL, hence, the term “two-point modulation.” In two-point modulation, a second modulation signal is inserted into the PLL after the loop filter. An example of a two-point phase modulator is shown in FIG. 1. The two-point phase modulator includes a phase frequency detector 25, a loop filter 65 (which is a low-pass (LP) filter), an adder 11, a voltage controlled oscillator (VCO) 16, a frequency divider 8 in the feedback loop, and a sigma delta modulator 9. A post PLL power amplifier 14 is also present for adding the amplitude part. A similar modulation scheme is described in U.S. Pat. No. 5,834,987, entitled “Frequency synthesizer systems and methods for three point modulation with a DC-response,” which is incorporated herein by reference.
In operation, the instantaneous frequency finst of the baseband signal is applied to the PLL 15 at two points: point 10 (at the sigma delta modulator) and point 12 (at the adder). A reference frequency θref is applied to the phase frequency detector 25, and an amplitude part “A” is applied to the power amplifier 14. The transfer function from the modulation inputs to the output of the VCO 16 can be derived as:
                                                                                          θ                                      out                    ,                    VCO                                                  ⁡                                  (                  s                  )                                            =                            ⁢                                                                                                                  f                        inst                                            ⁡                                              (                        s                        )                                                                                    N                      ⁢                                                                                          ⁢                      s                                                        ⁢                                                                                    K                        phd                                            ⁢                                                                        K                          vco                                                s                                            ⁢                                                                                          ⁢                                                                        H                          LP                                                ⁡                                                  (                          s                          )                                                                                                            1                      +                                                                        K                          phd                                                ⁢                                                                              K                            vco                                                    s                                                ⁢                                                                                                            H                              LP                                                        ⁡                                                          (                              s                              )                                                                                N                                                                                                                    +                                                                                                      ⁢                                                                                          f                      inst                                        ⁡                                          (                      s                      )                                                                            K                    vco                    ′                                                  ⁢                                                                                                    K                        vco                                            s                                        ⁢                                                                                                                      1                    +                                                                  K                        phd                                            ⁢                                                                        K                          vco                                                s                                            ⁢                                                                                                    H                            LP                                                    ⁡                                                      (                            s                            )                                                                          N                                                                                                                                                                    =                            ⁢                                                                                          f                      inst                                        ⁡                                          (                      s                      )                                                        s                                ⁢                                                                                                    K                        vco                                                                    K                        vco                        ′                                                              +                                                                  K                        phd                                            ⁢                                                                        K                          vco                                                s                                            ⁢                                                                                                    H                            LP                                                    ⁡                                                      (                            s                            )                                                                          N                                                                                                  1                    +                                                                  K                        phd                                            ⁢                                                                        K                          vco                                                s                                            ⁢                                                                                                    H                            LP                                                    ⁡                                                      (                            s                            )                                                                          N                                                                                                                                                                    =                            ⁢                                                [                                      If                    ,                                                                  K                        vco                                            =                                              K                        vco                        ′                                                                              ]                                =                                                                            f                      inst                                        ⁡                                          (                      s                      )                                                        s                                                                                        (        1        )            
As can be seen, the transfer function for the two-point modulator is independent of the PLL loop bandwidth. This eliminates the trade-off between PLL loop bandwidth and modulation bandwidth. Unfortunately, because the transfer function is dependent on the VCO gain, Kvco, the scheme results in a new unknown being introduced, namely, the estimation of the VCO gain, K′vco. If K′vco is wrong, then spectral growth may result that may compromise the ACPR (adjacent channel power ratio) requirement of the system.
A standard VCO configuration is depicted in FIG. 2. As can be seen, the VCO includes a resonator composed of inductors L1, L2 (20, 22) and varactors Cv (30, 32). Parasitic capacitance Cpar (24) represents all capacitor loading and all parasitic capacitances as seen from the resonator. Also present is a tuning network composed of coupling capacitors Cc (26, 28) and Rgnd (34, 36) (ground reference for the varactors) for coupling the varactors Cv loosely to the resonator. The bottom part of FIG. 2 shows the active components (e.g., transistors 38, 40) responsible for sustaining the oscillation. In a radio frequency (RF) ASIC with an onboard VCO, the VCO gain is dependent on the size of the inductor, the output frequency, and the bias point of the varactor.
The tuning sensitivity (VCO gain) of the VCO is derived by taking the derivative of the VCO center frequency ωo with respect to the tuning voltage, as follows:
                                                                                                              w                    o                                    =                                      1                                                                                            L                          tot                                                ⁢                                                  C                          tot                                                                                                                    ;                            ⁢                                                                                                                                                                                                                          ∂                                                  w                          o                                                                                            ∂                                                  V                          tune                                                                                      =                                        ⁢                                                                                            ∂                                                      w                            o                                                                                                    ∂                                                      C                            tot                                                                                              ⁢                                                                        ∂                                                      C                            tot                                                                                                    ∂                                                      C                            v                                                                                              ⁢                                                                        ∂                                                      C                            v                                                                                                    ∂                                                      V                            tune                                                                                                                                                                                                                =                                        ⁢                                                                  -                                                                              L                            tot                                                                                2                            ⁢                                                                                          (                                                                                                      L                                    tot                                                                    ⁢                                                                      C                                    tot                                                                                                  )                                                                                            3                                2                                                                                                                                                        ⁢                                              1                        2                                            ⁢                                                                        (                                                                                    C                              c                                                                                                                      C                                c                                                            +                                                              C                                V                                                                                                              )                                                2                                            ⁢                                                                        ∂                                                      C                            V                                                                                                    ∂                                                      V                            tune                                                                                                                                                                                                                =                                        ⁢                                                                  -                                                                                                            L                              tot                                                        ⁢                                                          w                              o                              3                                                                                2                                                                    ⁢                                              1                        2                                            ⁢                                                                        (                                                                                    C                              c                                                                                                                      C                                c                                                            +                                                              C                                V                                                                                                              )                                                2                                            ⁢                                                                        ∂                                                      C                            V                                                                                                    ∂                                                      V                            tune                                                                                                                                                                                                      (        2        )            
As can be seen from Equation (2), the tuning sensitivity is dependent on many parameters. For example, the VCO on-chip inductors (e.g., L1, L2) is a large metal structure and is inherently stable. The varactor capacitance and the slope of the varactor capacitance are dependent on the tuning voltage Vtune (42). The tuning voltage Vtune, in turn, is dependent on the VCO center frequency. By making a careful design and keeping the above equation in mind, however, the total VCO gain variation can be reduced.
A table with measured VCO gain versus frequency can compensate for variations in the VCO gain. The main problem with this solution, however, is that when manufacturing the circuits, the parasitic capacitance (Cpar) of the resonator varies, and therefore a different tuning voltage is required to get the correct output frequency. The VCO gain may vary as much as 50% from one sample to another. This means that the VCO gain would have to be measured for each VCO chip to get stable performance.
An alternative solution is described in U.S. Pat. No. 5,834,987, which is a modified VCO circuit configuration where the VCO has two separate inputs, one for the PLL tuning voltage and one for the modulation input. This type of circuit configuration is depicted in FIG. 3. As can be seen, the circuit of FIG. 3 is similar to the circuit of FIG. 2 except that a separate tuning input Vmod (50) and modulation varactors CV1 (30-1, 32-1) are added for modulation. Coupling capacitors CC1 (26-1, 28-1), and grounding resistors Rgnd1 (34-1, 36-1) are also present. The Vmod tuning input is similar to the Vtune tuning input (42), but has a DC voltage applied to set the operating point of the varactors CV1. This allows the modulation varactors CV1 to be biased at a suitable DC level. Also, the input bandwidth and tuning sensitivity can be optimized for modulation. If the DC level applied to the varactors CV1 is constant, the only thing varying in Equation (2) is the center frequency. In other words, the modified VCO solution is independent of parasitic capacitor variations, since such variations are compensated in the tuning voltage. This means that the VCO gain variation from sample to sample is mainly dependent on spread in the varactor at the specific bias point and spread in the coupling capacitor. But by careful design, the VCO gain variation can be made less than 10% (mainly by choosing large size components).
Although the above described designs have merit, they may not be sufficient for some systems with strict requirements for VCO gain estimation, such as EDGE and WCDMA systems. Moreover, for future systems with more complex modulation schemes (e.g., 16QAM), the requirement of VCO gain estimations will be even higher. Therefore, some kind of automatic calibration or trimming of the VCO gain is needed.
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