1. Field of the Invention
The present invention relates to a demodulation system, and more particularly to a demodulation system including an automatic equalizer for use on the reception side of a digital radio communication system using a multilevel quadrature amplitude modulation (multilevel QAM) method or a polyphase modulation method.
2. Description of the Related Art
In recent years, digital radio communication systems have employed, on the reception side thereof, demodulation systems equipped with equalizers that use transversal filters for preventing deterioration in the signal quality due to frequency selective fading produced in propagation paths.
In particular, in a demodulation system in which a decision feedback equalizer is used as the equalizer, and moreover, an adaptive matched filter precedes the decision feedback equalizer, even within a range in which .rho.&gt;1 (.rho. is the ratio of the amplitude of a reflected wave to the amplitude of a principal wave) where sufficient equalizing capability cannot be obtained solely through the use of a decision feedback equalizer, the same equalizing capabilities can be obtained as for the range 0&lt;.rho.&lt;1. Fading in the range .rho.&gt;1 is fading in which an interference wave exists at a position advanced in time in relation to the principal signal and is referred to as non-minimum phase shift type fading. Fading in the range 0&lt;.rho.&lt;1 is fading in which an interference wave exists at a position delayed in time in relation to the principal signal, and is referred to as minimum phase shift type fading. This type of demodulation system is disclosed in an invention by the same inventor as the present invention, Japanese Patent Application Laid-open 92-271508, being the earlier application of U.S. Pat. application Ser. No. 07/842,422, which has issued as U.S. Pat. No. 5,321,723.
FIG. 1 shows the construction of a demodulation system of the prior art described above. An intermediate frequency band modulated signal Si is supplied to the input terminal 1. The input terminal 1 is connected to a demodulator 11. The demodulator 11 demodulates the modulated signal Si and outputs an analog baseband signal AB. The analog baseband signal AB is supplied to an analog-digital (A/D) converter 12. A clock signal CLK is also supplied to the A/D converter 12 from the demodulator 11. The A/D converter 12 uses the clock signal CLK to sample-quantize the analog-baseband signal AB and output an M-bit parallel digital signal Sd. This M-bit parallel digital signal Sd is supplied to an adaptive matched filter 13. A polarity signal A is included within the M-bit signal Sd. A polarity signal D is included within the signal outputted by the adaptive matched filter 13. The adaptive matched filter 13 includes a transversal filter (not shown). The adaptive matched filter 13, by controlling the transversal filter within it by tap coefficients which were generated by time-averaging and correlating the polarity signal A and the polarity signal D, can make symmetrical the impulse response of the propagation path. As a result, non-minimum phase shift type fading as well as minimum phase shift type fading can be split into two types of fading having substantially equal interference before and after in time relative to the principal signal. In this case, the amount of interference in each of the two types of fading split in this manner is less than the amount of interference of fading of a signal inputted to the adaptive matched filter 13. In any case, the adaptive matched filter 13 outputs as its output signal a matched signal Sm including the above-described polarity signal D.
Matched signal Sm is supplied to a decision feed-back equalizer 14. An error signal E is included within an output signal from the decision feedback equalizer 14. This error signal E indicates the polarity of divergence from the predetermined value of post-equalizing signal Se. The decision feedback equalizer 14 includes a decision feedback transversal filter (to be described hereinafter). The decision feedback equalizer 14, by controlling the decision feedback transversal filter within it by tap coefficients obtained by time-averaging and correlating the polarity signal D and the error signal E, can eliminate intersymbol interference that occured in fading.
FIG. 2 shows the detailed structure of the decision feedback equalizer 14. The decision feedback equalizer 14 has a decision feedback transversal filter 31, and a control signal generating circuit 32. The transversal filter 31 is composed of a pre-equalizer 41, a post-equalizer 42, an adder 43, and a decision circuit 44. The control signal generating circuit 32 has a control signal generating circuit 45 for the pre-equalizer and a control signal generating circuit 46 for the post-equalizer. The pre-equalizer 41 has the capability to eliminate interference waves advanced in time relative to the principal signal, and the post-equalizer 42 has the capability of eliminating interference waves that were delayed in time relative to the principal signal. In particular, because the post-equalizer 42 takes as its input signal the equalized decision signal SD and eliminates interference waves based on this decision signal SD, virtually all interference can be eliminated.
FIG. 3 shows equalizing characteristics for two-ray fading of a decision feedback equalizer with the adaptive matched filter described above. FIG. 3 shows what is referred to as signature characteristics, the horizontal axis showing notch position .DELTA.fd which is the shift of fading notch frequency from the center of the modulation spectrum of a desired signal, the shift being normalized by the clock frequency, and the vertical axis showing the amplitude ratio .rho. which is the amplitude of the reflected wave (delayed wave) normalized by the amplitude of the principal wave. The notch depth Dn is expressed by Dn=-20 log (1-.rho.)dB. Using notch position fd and amplitude ratio .rho. as parameters, curve S is plotted by interconnecting points fd and .rho. where error ratio P=1.times.10.sup.-4. Error ratio P is greater than 10.sup.-4 in the area that is surrounded by curve S in FIG. 3. Therefore, it can be understood that the smaller the area surrounded by curve S, the greater the ability of the equalizer. The signature curve S shown in FIG. 3 shows that equalizing is possible outside the vicinity of .rho.=1.
FIG. 4 shows the signature characteristics for a demodulation system equipped with only a decision feedback equalizer that does not use an adaptive matched filter. When 0&lt;.rho.&lt;1, interference waves exist only in delayed positions in time relative to the principal signal, and consequently, the post-equalizer 42 of the decision feedback equalizer 14 has sufficient effect to carry out complete equalizing. Because interference waves exist at advanced positions in time relative to the principal signal when .rho.&gt;1, the pre-equalizer 41 of the decision feedback equalizer 14 operates. However, because the pre-equalizer 41 is inputting signals distorted by fading, it lacks sufficient equalizing effect, and the signature characteristics when .rho.&gt;1 are poor. When an adaptive matched filter is added to the decision feedback equalizer, the signature characteristics for .rho.&gt;1 are improved as shown in FIG. 3.
Although a demodulation system of the prior art which is preceded by an adaptive matched filter as explained above has remarkably improved characteristics for fading in a range where .rho.&gt;1 as compared with a demodulation system equipped with only a decision feedback equalizer, such a system is problematic in that, for fading in a range where 0&lt;.rho.&lt;1, it has inferior signature characteristics compared with a demodulation system equipped with only a decision feedback equalizer.
USPTO Backgrounds
USPTO Backgrounds