In many mobile radio systems such as GSM (Global System for Mobile Communications) and its further development EDGE (Enhanced Data Services for GSM Evolution), the overall transmission bandwidth is subdivided into a large number of narrowband subscriber frequency bands (traffic channels). The bandwidth of a subscriber frequency band in GSM and EDGE systems is 200 kHz. FIGS. 1a to 1c show three important interference influences during reception of a narrowband payload signal such as this.
FIGS. 1a to 1c respectively show the spectral profile of a received signal 1 in the presence of interference 2.1, 2.2 and 2.3. FIG. 1a shows the narrowband received signal 1 in the presence of noise, which represents broadband interference 2.1. FIGS. 1b and 1c show two cases of multiple access interference, of multiple access interference (MAI), specifically cochannel interference (FIG. 1b) and adjacent channel interference (FIG. 1c). While the interference 2.2 in the case of cochannel interference occurs in the same subscriber frequency band as the desired signal 1, and is caused, for example, by an active subscriber in another cell in the network, the interference 2.3 in the case of adjacent channel interference occurs in one of the two adjacent subscriber frequency bands.
The influence of adjacent channel interference is influenced by the channel width of the subscriber frequency bands and the symbol frequency used in the system. In order to achieve a high system subscriber capacity and a high data rate, it is desirable to use narrow channel widths and high symbol frequencies. On the other hand, this results in an increase in the adjacent channel interference which, however, must not exceed a specific limit.
In the case of GSM and EDGE, the symbol frequency is 270.833 kHz and the channel width, as already mentioned, is 200 kHz. This means that the desired signal 1 and the interference 2.3 caused by adjacent channel interference spectrally overlap one another, as is shown in FIG. 1c. It is not possible to completely suppress the adjacent channel interference without constraining the spectrum of the desired signal 1.
In conventional receivers for mobile communications systems, the channel filter which is used to filter out the desired subscriber frequency band has a fixed, predetermined bandwidth. The chosen bandwidth represents a compromise between the mutually contradictory aims of utilization of the subscriber frequency band as well as possible for signal detection and suppression of adjacent channel interference as well as possible. This compromise is necessarily sub-optimal in many receiving situations.
The German Patent Application DE 101 52 628.8, which only represents the prior art in accordance with §3(2) of the German Patent Act with reference to the present application, has proposed an adaptive channel filter for mobile radio receivers and a method for adaptive channel filtering, in which the pass bandwidth of the channel filter is set as a function of the strength of the adjacent channel interference. This results in an adaptive channel filter by means of which the payload signal can always be optimally filtered in different receiving and interference situations.
FIG. 2 shows an embodiment which was described in this older application. The adaptive channel filter has a filter 200 with a variable pass bandwidth, and has a control device 30 for setting the pass bandwidth of the filter 200. The adaptive channel filter is preferably a digital low-pass filter, which is located in the baseband processing section of a mobile radio receiver. The signal 40 which is supplied to the adaptive channel filter has, for example, already been frequency-selected and/or subscriber-selected by suitable down-mixing of the frequency of the desired subscriber frequency band to baseband, but has not yet or has not been adequately bandwidth-limited.
The filter 200 which is shown with the dashed boundary and has a variable pass bandwidth has a first low-pass filter 200.2 which has a cut-off frequency above the desired signal. The filter 200 also has a series arrangement of the low-pass filter 200.2 and of a downstream constriction or limiting filter 200.3. The constriction filter 200.3 has the function of somewhat reducing the spectral pass band of the low-pass filter 200.2, that is to say the series arrangement of the filters 200.2 and 200.3 behaves like a single low-pass filter with a cut-off frequency which is lower than the cut-off frequency of the low-pass filter 200.2.
The outputs of the low-pass filters 200.2 and 200.3 are passed to the inputs of a selection switch 210. The selection switch 210 has a control input 22, via which one of the supplied filter signals can be selected and can be switched with a variable pass bandwidth to an output 23 of the filter 200.
A bandpass filter 200.4, to which the signal 40 is likewise supplied, is connected in parallel with the channel filter 200 with the variable pass bandwidth. The bandpass filter selects the spectral component from the adjacent channel interference source from the signal 40. The principle of operation of the adaptive channel filter shown in FIG. 2 is based on a power comparison between the signals x, and x2, which are filtered by the bandpass filter 200.4 and the low-pass filter 200.2 and are supplied to the control device 30. When strong adjacent channel interference is present, the power of the signal which is emitted from the low-pass filter 200.2 is relatively small in comparison to the power of the signal which is filtered by the bandpass filter 200.4, since the bandpass filter 200.4 passes a higher interference power than the low-pass filter 200.2. If the ratio of the two signal power levels exceeds a threshold value which is defined by the user, the selection switch 210 is actuated by the control device 30 such that the output of the low-pass filter 200.3, and thus of the series circuit comprising the low-pass filters 200.2 and 200.3, is produced with the lower overall cut-off frequency at the output 23 of the filter 200. Conversely, that is to say if the adjacent channel interference is low or is negligible, the ratio of the two power levels is below the predetermined threshold value, in which case the output of the low-pass filter 200.2 is selected by the selection switch 210, and is passed to the output 23. The bandpass filter 200.4 can be designed such that it extracts precisely that part of the signal power which is most valid for a power comparison in the control device 30.
The complex sample values x1(k) and x2(k) which are calculated by the filters 200.4 and 200.2 are passed to the control device 30. In each signal path, the control device 30 has an energy estimator 31 or 32, respectively, each of which contains a magnitude forming device and an accumulator, in this sequence. The energy estimator 31 in the path which is associated with the sample values x1(k) is followed by a multiplier 33, which multiplies the sample values by a threshold value preset value t which can be defined by the user. The output of the multiplier 33 and the output of the energy estimator 32 in the other path are supplied to the two inputs of a comparator 34. The comparator 34 checks which of the two inputs has the greater value, and produces a corresponding comparison signal at its output. This is supplied in the manner which has already been described as a control signal to the input 22 of the selection switch 210.
In the energy estimators 31 and 32, the magnitude forming devices and the accumulators in each case calculate the sum of the magnitudes of the real and imaginary parts of both input signals over the accumulation time period which, for example, is the duration of a burst. This results in the adaptive channel filter having a behaviour which is adapted on a burst basis. The equation for the calculation of the output variables P1 and P2 is:
                              P          i                =                              ∑                                          k                =                1                            ,              …              ,              K                                ⁢                                    (                                                                                      Re                    ⁡                                          (                                                                        x                          i                                                ⁡                                                  (                          k                          )                                                                    )                                                                                        +                                                                        Im                    ⁡                                          (                                                                        x                          i                                                ⁡                                                  (                          k                          )                                                                    )                                                                                                    )                        ⁢                          (                                                i                  =                  1                                ,                …                ⁢                                                                  ,                N                            )                                                          (        1        )            
where N is the number of inputs of the control device 30, x;(k) are the sample values with the time index k supplied to the i-th input of the control device 30, and K is the number of sample values in a burst.
Instead of forming the sum of the magnitudes of the real and imaginary parts on the input signal, it is also possible to add the squares of the magnitudes.
The variables P1 and P2 are used as estimates of the respective signal power levels. The multiplier 33 multiplies the variable P1 by the threshold value preset value t. The variable PjXt is compared with the variable P2 in the comparator 34.
The adaptive channel filter which is illustrated in FIG. 1 has been implemented in GSM and EDGE receivers. An IIR (Infinite Impulse Response) filter with nine coefficients has been used for the bandpass filter 200.4. The low-pass filter 200.2 with a high cut-off frequency has been configured as a linear FIR (Finite Impulse Response) phase filter with 33 symmetrical coefficients. The constriction filter 200.3 has been chosen as a linear FIR phase filter with 13 symmetrical coefficients. The oversampling used in the receiver was m=2.
Thus, overall, a channel filter with a wide pass bandwidth is chosen when the adjacent channel interference is low, and a channel filter with a narrow pass bandwidth is used when the adjacent channel interference is high, and its desired frequency response is produced by cascading the low-pass filter 200.2 with a high cut-off frequency and the constriction filter 200.3. The ratio of the energy in the payload signal to the energy from the adjacent channel interference source is used as the criterion for selection of the channel filter. In this case, the energy from the adjacent channel interference source is multiplied by a predefined threshold t, and is compared with the energy from the adjacent channel interference source. If Pjt is less than P2, the output of the low-pass filter 200.2 is taken, otherwise the output from the constriction filter 200.3 is used.
FIG. 1 is based on the assumption that the clock rate of the signal is 40 m×fT. The symbol frequency is denoted fT, and is 270.833 kHz for GSM and EDGE. The oversampling factor is denoted m. For a channel filter in baseband, m is typically equal to 2. FIG. 1 shows that, with oversampling using the factor m, optional signal decimation can be carried out in each case between the low-pass filters 200.2 and 200.3 and the selection switch 210. The (optional) decimation is carried out by the decimators 211. The method of operation of each decimator 211 is to pass onto the output only one sample value from a group of m sample values, with the remaining m−1 sample values being rejected. The signal decimation is required only when signal processing is carried out at the symbol clock rate downstream from the adaptive channel filter.
The adaptive channel filter in FIG. 1 has the following disadvantage, however.
Owning to non-linearities in the RF receiver, each signal component (the payload signal or interference signal) at the input of the RF receiver also leads to a corresponding DC component (direct current, DC offset) in the quadrature-demodulated I and Q signals at the output, as is shown in FIG. 2a. In certain receiving situations, the DC offset can also vary within one burst. In this case, interference in the form of a step, a “DC step” is superimposed on the I and Q output signals, as is illustrated by way of example in FIG. 2b. The superimposition of a DC offset or DC step in the I and Q signals corrupts the estimate of the payload signal energy, while the DC interference in the energy estimate for the adjacent channel interference is largely suppressed owing to the bandpass filtering. Residual interference admittedly remains in the transition area of the DC step even after the bandpass filtering, but this lasts for a negligible time in comparison to the burst duration for energy estimation. The corruption of the energy measurement in one of the two paths leads to an increased error rate in the detection of the adjacent channel interference source, and thus to a deterioration in the reception quality.
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