Known types of tuners for receiving terrestrially broadcast television signals generally use the single conversion architecture in order to select a desired channel, for example from a received frequency spectrum from 50 to 860 MHz, and to convert this to an output intermediate frequency, for example of 44 MHz. In order to “protect” the desired channel from potentially interfering signals such as the image channel, such tuners typically contain a plurality of tracking filters. Such filters typically comprise or include tracking bandpass filters with a passband center frequency centered on the desired channel. The presence of such filters attenuates many of the undesired channels before a single frequency converter so as to reduce the effects of intermodulation.
In order to cover the whole broadcast spectrum, it is known for such a tuner to comprise three sub-tuners 1, 2 and 3 as shown in FIG. 1 of the accompanying drawings. The sub-tuners 1, 2 and 3 have a common radio frequency (RF) input 4 and a common intermediate frequency (IF) output 5. The tuners 1, 2 and 3 are used for reception in a low band (50 to 170 MHz), a mid band (170 to 440 MHz) and a high band (440 to 860 MHz), respectively. The sub-tuners are of the same construction, a typical example of which is illustrated in FIG. 2 of the accompanying drawings.
The RF input 10 is connected to a first tuneable bandpass filter 11. The filter 11 is of the “single element” type and comprises a single inductor/capacitor resonant network whose center frequency is arranged to track with the frequency of the desired channel which has been selected for reception. The filter 11 thus “selects” the desired channel from the full received frequency spectrum and provides a first attenuation to at least some of the undesired channels, including the image channel. The filter 11 therefore provides protection from intermodulation being generated in the immediately following stage.
In this example, the tuner uses high side mixing such that the frequency of a local oscillator (LO) 12 is above the frequency of the selected channel and differs therefrom by the output intermediate frequency. The image channel is therefore above the selected channel and spaced therefrom by twice the output intermediate frequency.
The output of the filter 11 is supplied to a low noise amplifier/automatic gain control (LNA/AGC) stage 13, which provides a first system variable gain. The output of the stage 13 is supplied to a further tuneable bandpass filter 14. The filter 14 is of dual element type and comprises two resonant networks generally arranged as a double-tuned loosely-coupled arrangement whose center frequency is arranged to track the desired channel frequency. The filter 14 provides further but higher Q attenuation to the undesired channels including the image channel.
The output of the filter 14 is supplied to a mixer 15 forming part of a frequency changer, which also comprises the local oscillator 12 controlled by a phase blocked loop (PLL) synthesizer 16. The IF output of the mixer is supplied via a roofing filter 17 and an amplifier 18 to the output 19 of the tuner. The roofing filter 17 reduces the composite power supplied to the amplifier 18 so as to prevent overload distortion effects.
The synthesizer 16 operates in the well-known way and controls the local oscillator frequency so as to convert a desired channel to the output intermediate frequency. The synthesizer 16 has a control voltage output 20, which is supplied to the local oscillator 12 and also to the frequency control inputs of the filters 11 and 14.
In a typical example of such a known tuner arrangement, the mixer 15, the amplifier 18 and the local oscillator for all three sub-tuners 1, 2 and 3 together with a synthesizer 16 which is common to the three sub-tuners are disposed in a common integrated circuit. The tracking filters 11 and 14 and the stage 13 are formed on a separate substrate for each of the sub-bands and comprise a plurality of discrete components.
The tracking filters 11 and 14 and the local oscillator 12 generally include similar resonant networks formed from varactor diodes and air core inductors in the form of air coils. These networks are arranged such that their resonant frequencies substantially track over the required operating frequency range with a frequency offset equal to the intermediate frequency between the filter networks and the oscillator network. During production, the tracking alignment between the filters 11 and 14 and the oscillator 12 is adjusted for a best compromise across the required frequency range by manual adjustment of the air coils. This typically involves moving the coils closer together or further apart so as to adjust their inductance and hence the characteristic response at a plurality of different frequencies. It is thus possible to provide RF filtering ahead of the frequency changer 12, 15 capable of providing a tracking bandwidth of between 3 and 6 channels and an image cancellation or reduction of typically 55 dB.
A typical example of the single element filter 11 is shown in FIG. 3 of the accompanying drawings. The filter has an RF input 21 connected to a first inductor element 22, which is inductively coupled to a second inductor element 23. A padding and DC blocking capacitor 24 is disposed between ground and the inductor element 23. A varactor diode 25 is connected in parallel with the inductor element 23 to form a parallel resonant circuit. The resonant circuit is connected to the output 26 of the filter via a DC blocking capacitor 27. The capacitance of the varactor diode 25, and hence the resonant frequency of the parallel resonant circuit, is controlled by a control voltage Vvar supplied to the diode 25 via an isolating resistor 28.
Following assembly, the filter 11 is aligned during a manual or semi-automatic alignment step. In particular, the inductive coupling between the first and second elements 22 and 23 is adjusted and the inductances of the elements 22 and 23 is adjusted so as to optimize the coupling frequency range and so as to align the frequency versus voltage characteristic in order for the filter 11 to track optimally with the local oscillator.
FIG. 4 of the accompanying drawings illustrates an example of the dual element filter 14. The filter has an RF input 30 connected via a coupling capacitor 31 to a parallel resonant circuit comprising an inductor element 32, a padding and DC blocking capacitor 33 and a varactor diode 34. The element 32 is inductively coupled to an inductor element 35 forming part of a parallel resonant circuit comprising a padding and DC blocking capacitor 36 and another varactor diode 37. The second resonant circuit is connected to the output 38 of the filter via a coupling capacitor 39. The frequency control voltage Vvar is supplied to the diodes 34 and 37 via isolating resistors 40 and 41, respectively.
The filter 14 is also subjected to an alignment procedure towards the end of manufacture. Again, the coupling between the inductor elements 32 and 35 and the inductance values of the elements 32 and 35 are adjusted so as to optimize the frequency verses voltage characteristic for tracking with the local oscillator. Further, the coupling is optimized so as to maximize the passband flatness provided by the double-tuned resonant network.
For tuners of the type shown in FIG. 1, each of the sub-tuners contains examples of the filters 11 and 14 with each filter being optimized for the desired frequency range of operation. In practice, the values of the inductor elements in the filters may range from a few hundred nano Henries (nHs) in the low band sub-tuner 1 to a few nHs in the high band sub-tuner 3. These inductor elements are characterized by high Q factors, for example in excess of 50. The Q factors of the inductor elements and of the varactor diodes are important as they determine the ratio of passband to resonant frequency and hence the attenuation ratio provided between the center of the passband and offset frequencies.
Such filters 11 and 14 typically have a tuning range in excess of one octave. This is achieved through the capacitance range of the varactor diodes 25, 34 and 37, which typically provide a ratio of 12:1 between their maximum and minimum capacitances.
Such filters are not suitable for integration, for example in an integrated circuit. In particular, the inductance values for the low band are too large to be practically implemented. Also, the total composite inductance is too large to be practically implemented. Further, the Q factor of integrated circuit inductors is significantly lower than for air coils and integrated inductors cannot be manually adjusted so as to align more than one resonant network. Because of the planar nature of integrated circuits, it is not practical to form inductively coupled arrangements which can be manually or electronically adjusted. Integrated circuit varactor diodes have a significantly smaller capacitance ratio than discrete varactor diodes and cannot support the relatively high voltages which are typically required to achieve a sufficiently large tuning range. Also, integrated circuit varactors diodes have a lower Q factor than discrete varactor diodes and this, combined with the limited Q factor of integrated circuit inductors, limits the filtering performance which can be achieved. Thus, it is not practical to integrate fully a single conversion tuner of the type illustrated in FIGS. 1 to 4.
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