In the wireless communication systems for next-generation mobile communications, it is important to realize high-speed data transmission. However, as the data rate increases, interference between symbols due to multipaths, i.e. multipath interference, arises as a problem. There are various methods for suppressing this multipath interference. There is a linear equalizer as a relatively simple method and a frequency equalizer is proposed for carrying out this equalization process in frequency domain. For example, see prior art document 1 (D. Falconer, S. L. Ariyavisitakul, A. Benyamin-Seeyar, and B. Eidson, “Frequency Domain Equalization for Single-Carrier Broadband Wireless Systems,” IEEE Commun. Mag., vol. 40, no. 4, pp. 58-66, April 2002.).
FIG. 1 shows one example of the configuration of a conventional equalizer in which a plural path samples method described in prior art document 2 (Matsumoto, Yoshida, and Ushirokawa, “Study of Accurate Channel Separation Method in MMSE Chip Equalizer for HSDPA Terminal” 2005 The Institute of Electronics, Information and Communication Engineers General Conference, B-5-120.) is applied to a frequency equalizer described in the above prior art document 1. The conventional equalizer comprises a receiving antenna 1, a path timing detecting section 2, a detected path transmission channel estimating section 3, an adjacent path transmission channel estimating section 4, a transmission channel response vector generating section 5, serial/parallel (S/P) converting sections 6 and 10, fast Fourier transform (FFT) sections 7 and 11, a weight calculating section 8, a guard interval (GI) removing section 9, an equalization filter 12, an inverse fast Fourier transform (IFFT) section 13, a parallel/serial (P/S) converting section 14, and a noise power estimating section 15.
The equalizer to which the plural path samples method is applied has a feature that each path is represented by a plurality of transmission channel estimation value samples, thereby improving the equalization characteristics in an environment where paths exist close to each other and providing tolerance to path timing errors.
The receiving antenna 1 receives a digitally modulated single-carrier signal. The path timing detecting section 2 receives as its input an oversampled received signal and detects the timings of a plurality of paths using pilot signals included in the received signal. As a method of detecting the timings, use is made of a method of detecting the timings of a plurality of paths having high levels based on the results of detection of sliding correlation between pilot signals included in the received signal and a known pilot signal sequence, or the like. The detected path transmission channel estimating section 3 receives as its input the oversampled received signal and the path timings detected by the path timing detecting section 2 and estimates transmission channel estimation values at the timings of the detected paths using the pilot signals included in the received signal.
The adjacent path transmission channel estimating section 4 receives as its input the oversampled received signal and the path timings detected by the path timing detecting section 2 and, using the pilot signals included in the received signal, estimates transmission channel estimation values at a plurality of timings (adjacent path timings) before and after each of the detected path timings. The transmission channel response vector generating section 5 receives as its input the transmission channel estimation values estimated by the detected path transmission channel estimating section 3 and the adjacent path transmission channel estimating section 4 and generates a transmission channel response vector.
FIG. 2 is a diagram showing the state of generation of a transmission channel response vector in the transmission channel response vector generating section 5. A solid line represents a transmission channel estimation value at a detected path timing and a broken line represents a transmission channel estimation value at an adjacent path timing. A transmission channel response vector is generated by concatenating the transmission channel estimation values at the detected path timings and the adjacent path timings (transmission channel estimation values of plural path samples).
The S/P converting section 6 performs S/P conversion of the transmission channel response vector generated by the transmission channel response vector generating section 5. The FFT section 7 receives as its input the transmission channel response vector converted by the S/P converting section 6 and outputs a transmission channel estimation value converted into frequency domain. The noise power estimating section 15 receives as its input the oversampled received signal and the transmission channel estimation values estimated by the detected path transmission channel estimating section 3 and estimates a noise power.
The weight calculating section 8 receives as its input the frequency-domain transmission channel estimation value being the output of the FFT section 7 and the noise power estimated by the noise power estimating section 15 and calculates a weight of the equalization filter by the minimum mean square error method (MMSE). Given that a transmission channel estimation value at a subcarrier f used for converting the transmission channel response vector into frequency domain in the FFT section 7 is H(f), a weight W(f) of the equalization filter is expressed by the following formula (1).
                              W          ⁡                      (            f            )                          =                              H            *                          (              f              )                                                                                                            H                  ⁡                                      (                    f                    )                                                                              2                        +                          N              0                                                          (        1        )            where * represents a complex conjugate and N0 represents a noise power estimated by the noise power estimating section 15.
The GI removing section 9 receives as its input the oversampled received signal and removes a portion, corresponding to GI, of the received signal. The S/P converting section 10 performs S/P conversion of the received signal with GI removed by the GI removing section 9. The FFT section 11 receives as its input the received signal converted by the S/P converting section 10 and converts it into frequency domain. The equalization filter 12 receives as its input the equalization weight calculated by the weight calculating section 8 and the received signal frequency-converted by the FFT section 11 and performs equalization of the received signal in frequency domain.
The IFFT section 13 receives as its input a frequency-domain equalized signal being an output of the equalization filter 12 and converts it into time domain using IFFT. The P/S converting section 14 performs P/S conversion of the signal converted into time domain and outputs a demodulated signal.
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